Optical signals received through conventional optical links are typically distorted by significant amounts of chromatic dispersion (CD) and polarization dependent impairments such as Polarization Mode Dispersion (PMD), polarization angle changes and polarization dependent loss (PDL). Chromatic dispersion (CD) on the order of 30,000 ps/nm, and polarization rotation transients at rates of 105 Hz are commonly encountered. Various methods and systems intended to address some of these limitations are known in the art.
Applicant's U.S. Pat. Nos. 7,023,601 issued Apr. 4, 2006 and 7,266,306 issued Sep. 4, 2007, and Applicant's co-pending U.S. patent application Ser. Nos. 10/262,944 filed Oct. 3, 2002 and 10/307,466 filed Dec. 2, 2002 teach methods and systems for electronically compensating chromatic dispersion and polarization effects in a transmitter. FIGS. 1 and 2 schematically illustrate representative optical transmitters capable of implementing these methods.
In the transmitters of FIGS. 1 and 2, the transmitter 2 receives an input data stream x(t), and generates a corresponding output optical signal 4 for transmission through an optical link 6 of a communications system. A receiver (not shown in FIGS. 1 and 2) at the opposite end of the link operates to detect the optical signal, and recover the data stream x(t).
As may be seen in FIG. 1a, the transmitter 2 comprises a complex driver 8 which receives an input data stream x(t) and outputs analog drive signals VA(t) and VB(t) 10, which are supplied to respective branches of a complex optical modulator 12. The complex optical modulator 12 operates to modulate a continuous wave (CW) optical carrier signal 14 in accordance with the analog drive signals VA(t) and VB(t), to generate the output optical signal 4, which may be a carrier-suppressed optical signal. For simplicity of description, the digital input signal x(t) is considered to be a serial bit stream, but an encoded symbol stream (such as Phase Shift Keying, PSK, and Quadrature Phase Shift Keying, QPSK, symbols) may also be processed using these same techniques.
In the transmitter of FIG. 1a, the complex driver 8 comprises a digital filter 16, which implements a mapping function to generate respective multi-bit In-phase and quadrature values I(n) and Q(n) of a desired envelope of the optical E-field of the optical signal 4. Preferably, the mapping function also implements a compensation operator C[ ] to electrically pre-compensate impairments of the optical link 6, such as dispersion and polarization effects. This functionality is described in detail in applicant's co-pending U.S. patent application Ser. Nos. 10/262,944 filed Oct. 3, 2002; 10/307,466 filed Dec. 2, 2002; 10/405,236 filed Apr. 3, 2003, and International Patent Application No. PCT/CA03/01044 filed Jul. 11, 2003.
Various known digital filter types may be used to implement the digital filter 16, such as, for example, a Random Access Memory Look-up Table (RAM LUT). Alternatively, the digital filter 16 may be implemented using Finite Impulse Response (FIR) filters, Infinite Impulse Response (IIR) filters, and Fast Fourier Transform (FFT filters). In either case, the digital filter 16 generates the multi-bit In-phase and quadrature signal component values I(n) and Q(n) at a sample rate which is about double the baud-rate of the input signal x(t). Thus, for example, for a baud rate of 10 Gbaud, the sample rate will be about 20 GHz.
FIG. 1b illustrates an embodiment of the digital filter 16 known from Applicant's co-pending U.S. patent application Ser. No. 10/262,944 filed Oct. 3, 2002. In this embodiment, the input data stream x(t) is supplied to a deserializer 18 (such as a shift register) which converts the serial data stream into an n-bit parallel input vector, which is input to a Random Access Memory Look-Up Table (RAM LUT) 20. The RAM LUT 20 is pre-loaded with values of I(n) and Q(n) which are computed in advance for each possible value of the input vector, based on the compensation operator C[ ]. The width of the deserializer 18 (and thus also the input vector and the RAM LUT 20) is determined based on the maximum anticipated dispersion of the link 6. In some embodiments, this width may be 64 or 128 bits. For higher speed systems, it may be desirable to extend the width beyond 128 bits, for example to 512 bits or 1024 bits, so as to enable compensation of large amounts of dispersion (e.g. 30000 ps/nm or more) at line rates exceeding 10 GBaud. If desired, the RAM LUT 20 may be divided into blocks (not shown) spanning a portion of the width of the input vector, and the respective outputs of each of the blocks combined to obtain the final values of I(n) and Q(n). With the embodiment of FIG. 1b, obtaining a sample rate 1/TS at the output of the RAM LUT 20 that is double the baud rate of the input data stream x(t), can be obtained by computing, for each possible value of the input vector, a pair of I(n) and Q(n) values. A first I(n), Q(n) value is computed for the input vector with a phase shift of zero, while the second I(n), Q(n) value is computed for the input vector with a phase shift of T/2, where T is the bit- or symbol-period of the input data stream x(t). With this arrangement, the first and second I(n), Q(n) values can be latched out of the RAM LUT 20 during each bit-period T of the input data stream x(t).
A non-linear compensator 22 (which may also be implemented as a RAM LUT) is used to adjust the value of each successive sample I(n) and Q(n), to compensate non-linear performance of the transmitter 2, as described in applicant's co-pending U.S. patent application Ser. No. 10/262,944, filed Oct. 3, 2002; and International Patent Application No. PCT/CA03/01044 filed Jul. 11, 2003. The non-linear compensator 22 may be implemented as a separate device cascaded with the digital filter 16, as shown in FIG. 1, or may be “embedded” within the digital filter 16 by applying the mapping function implemented by the non-linear compensator 22 to the digital filter 16.
Respective high-speed Digital-to-Analog Converters (DACs) 24a,24b are used to convert the multi-bit sample values VI(n) and VQ(n) output from the non-linear compensator 22 into corresponding analog signals VA(t) and VB(t). If desired, the analog signals VA(t) and VB(t) can be conditioned, for example by means of respective filters 26a,26b and low noise amplifiers (LNA) 28a,28b, in a conventional manner, to remove out-of-band noise and to scale the signal amplitude to the dynamic range of the complex modulator 12.
As may be appreciated, the effects of the independent DACs 24a,24b, the filters 26a,26b and the LNAs 28a,28b for each signal may cause differential propagation delays between the non-linear compensator 22 and the optical modulator 12. Such differential delay can be compensated by means of digital filters 30a,30b located in at least one of the signal paths. Each digital filter 30a,30b can be controlled in a known manner to impose a selected delay, which is calculated to compensate for the differential propagation delays experienced by each of the signal components.
Referring now to FIG. 2, there is shown an embodiment of a system 32 which generates a polarization multiplexed optical signal 4′, in which respective different data streams xA(t) and xB(t) are modulated onto respective orthogonal transmitted polarizations 34 of the optical signal 4′. The system of FIG. 2 generally incorporates a pair of parallel transmitters 2 of the type shown in FIGS. 1a and 1b. In this case, each transmitter 2 receives a respective input signal xA(t) and xB(t), which may be independent data streams or may be derived from a single data stream. A common narrow band laser may be used for both complex modulators 12, as shown in FIG. 2, although separate lasers may also be used if desired. In either case, both complex modulators 12 operate at the same CW signal wavelength, and orthogonal polarizations.
The polarization multiplexed communications signal 4′ is generated by combining the respective optical signals 34a, 34b from each transmitter 2, using a polarization combiner 36. Respective polarization rotators 38a, 38b ensure orthogonal polarization states of the two optical signals 34a, 34b. This ensures that the two optical signals 34a, 34b are fully orthogonal, and thus can be combined into the polarization-multiplexed communications signal 4′ without interference.
Applicant's co-pending U.S. patent application Ser. Nos. 11/294,613 filed Dec. 6, 2005 and entitled “Polarization Compensation In A Coherent Optical Receiver”; 11/366,392 filed Mar. 2, 2006 and entitled “Carrier Recovery In A Coherent Optical Receiver”; and 11/423,822 filed Jun. 13, 2006 and entitled “Signal Acquisition In A Coherent Optical Receiver”, the content of all of which are hereby incorporated herein by reference, teach methods and systems for electronically compensating chromatic dispersion and polarization effects in a receiver. FIG. 3 schematically illustrates a representative coherent optical receiver capable of implementing these methods.
As may be seen in FIG. 3, an inbound optical signal 4″ is received through the optical link 6, split into orthogonal received polarizations X,Y by a Polarization Beam Splitter 40, and then mixed with a Local Oscillator (LO) signal 42 by a conventional 90° optical hybrid 44. The composite optical signals emerging from the optical hybrid 44 are supplied to respective photodetectors 46, which generate corresponding analog electrical signals. The photodetector signals are sampled by respective Analog-to-Digital (ND) converters 48 to yield raw multi-bit digital signals IX, QX and IY, QY corresponding to In-phase (I) and Quadrature (Q) components of each of the received polarizations.
Preferably, the raw multi-bit digital signals have resolution of n=5 or 6 bits which has been found to provide satisfactory performance at an acceptable cost. In the above-noted U.S. patent applications, the sample rate of the ND converters 48 is selected to satisfy the Nyquist criterion for the highest anticipated symbol rate of the received optical signal. Thus, for example, in the case of an optical network link 6 having a line rate of 10 GBaud, the sample rate of the A/D converters 48 will be approximately 20 GHz.
From the A/D converters 48, the respective n-bit signals IX, QX and IY, QY of each received polarization are supplied to a respective dispersion compensator 50, which operates on the raw digital signals to at least partially compensate chromatic dispersion of the received optical signal. The dispersion compensators 50 may be configured to operate as described in Applicant's co-pending U.S. patent application Ser. No. 11/550,042 filed Oct. 17, 2006.
The dispersion-compensated digital signals 52 appearing at the output of the dispersion compensators 14 are then supplied to a polarization compensator 54 which operates to compensate polarization effects, and thereby de-convolve transmitted symbols from the complex signals 52 output from the dispersion compensators 50. If desired, the polarization compensator 54 may operate as described in Applicant's co-pending U.S. patent application Ser. No. 11/294,613 filed Dec. 6, 2005 and Ser. No. 11/366,392 filed Mar. 2, 2006. The output of the polarization compensator 54 is a pair of multi-bit estimates X′(n) and Y′(n), 56 of the symbols encoded on each transmitted polarization 34 (FIG. 2). The symbol estimates X′(n), Y′(n) appearing at the output of the polarization compensator 54 are then supplied to a carrier recovery block 58 for LO frequency control, symbol detection and data recovery, such as described in Applicant's co-pending U.S. patent application Ser. No. 11/366,392 filed Mar. 2, 2006.
In the above described system, the dispersion compensators 50 operate across a large number of successive samples (e.g. 128 samples), which permits compensation of relatively severe chromatic dispersion, but at a cost of a relatively slow response to changing dispersion. This slow response is acceptable, because of the known slow rate of change of dispersion in real-world optical links. The polarization compensator 54, in contrast, is comparatively very narrow (e.g. on the order of about 5 samples), to enable a rapid update frequency, which is necessary to track observed high-speed polarization transients.
The above-described systems provide reliable signal acquisition, compensation of dispersion and polarization effects, carrier recovery and data recovery even in the presence of moderate-to-severe optical impairments. This, in turn, enables the deployment of a coherent optical receiver in real-world optical networks, with highly attractive signal reach and line rate characteristics. For example, a transmitter implementing the techniques described above with reference to FIG. 1 has demonstrated a signal reach of over 3000 km at a line rate of 10 GBaud, while a receiver implementing the methods described above with reference to FIG. 3 has demonstrated equivalent performance. It is noteworthy that this performance has been measured with real-time continuous processing, not just burst data acquisition followed by off-line processing or simulation. The receiver system described above with reference to FIG. 3 is the only coherent optical receiver known to the applicants to have achieved such real-time performance at multiple gigabaud.
A critical part of the design of an electronic dispersion compensation system, such as those described above with reference to FIGS. 1-3, is the sampling rate of the Digital-to-Analog converter (DAC) 24 in the transmitter and/or the Analog-to-Digital (ND) converter 48 in the receiver. It is standard practice to sample at an integer multiple (N) of the symbol rate. When N=1, the samples need to be aligned at the center of the eye in order that the signal can be accurately decoded. Because the signal bandwidth is greater than half of the sampling rate there will be a large amount of aliasing with N=1, precluding the application of frequency dependent digital filtering operations such as compensation for chromatic dispersion or Polarization Mode Dispersion (PMD).
Using the example of chromatic dispersion, the amount of phase shift that is caused by dispersion is proportional to the square of the frequency, and so is the phase shift of the compensation function. When some energy is aliased to appear at a frequency that is not the actual optical transmission frequency, then the wrong amount of dispersion compensation is applied to that energy. This energy then corrupts the eye of the received signal.
When these frequency dependent filtering operations are desired it is standard practice to sample at N≧2, so as to avoid aliasing. Thus, for example, in the case of an optical network link having a symbol rate of 10 GBaud, when N=2 the sample rate of the DAC and/or ND converters will be approximately 20 GHz.
More particularly, consider a system in which a baseband optical signal having a line (or symbol) rate of 1/T=10 Gbaud and a Bessel approximation to the ideal raised cosine spectrum (α=1.0), is sampled at a sample rate of 1/Ts=10 GHz (that is, N=Ts/T=1). This scenario, in which the sample period Ts=T, may be referred to as T-spaced sampling. Each sample of the sample stream is an impulse, so the frequency-domain spectrum of the sample stream will span a frequency range of 0-10 GHz, which encompasses the upper side-band of the optical signal, as may be seen in FIG. 4a. In this drawing, the lower side-band, spanning the frequency range between 0 and −10 GHz, is shown in dotted line. In fact, duplicates of the entire spectrum will repeat at 1/Ts=10 GHz intervals to infinity, as shown by the dashed lines, in FIG. 4b. As a result, the frequency-domain spectrum of the sample stream, between 0-10 Ghz will contain both the upper side-band of the baseband signal, and the lower side-band of the first order harmonic. The overlap between the baseband spectrum (centered at 0) and the first order harmonics (centered at ±10 GHz) represents aliasing. The use of a low-pass filter, as shown in FIG. 4c, to suppress frequencies above the base band reduces the aliasing (shaded regions of FIG. 4b), but not enough to prevent severe signal distortions. In this respect, it should be noted that in order to avoid severe effects upon the eye from phase distortion, the low pass filter normally used for this operation is an analog filter having a fifth order Bessel filter response, or an approximation to that shape. On the other hand, doubling the sample rate to 1/Ts=20 GHz (that is, T/2 or Nyquist sampling) causes the duplicate spectra to repeat at 1/Ts=20 GHz intervals. As may be seen in FIG. 4d, T/2 sampling eliminates the overlap between the baseband and 1st harmonic spectra, and thus distortions due to aliasing.
Professor Joseph Kahn of Stanford University stated at the IEEE LEOS Summer Topical Workshop in Portland Oreg., July 2007, that N could be as small as 3/2, and that N needed to be an integer multiple of ½ in order for the digital signal processing to be feasible. However, operation at N=3/2 would require an analog low pass filter with a very steep roll-off to suppress aliasing. For the purposes of the present disclosure, a “steep” (or, equivalently, a “sharp”) roll-off is considered to be a roll-off of greater than 20 dB per decade, for example 80 dB per decade. The desirable corner frequency is generally 1/2T, but can vary from that value with other design considerations and component tolerances. FIG. 5a illustrates the filter characteristic of a Chebychev filter having a suitable roll-off and a corner frequency of 17.5 GHz. As may be seen in FIG. 5b, such a filter inherently exhibits a highly non-linear group delay characteristic, as a function of frequency. This non-linear group delay characteristic causes unacceptable phase distortions to the received eye.
With increasing demand for link band-width, it would be desirable to increase the line rate beyond 1/T=10 Gbaud. For example, lines rates of 35 GBaud and higher have been proposed. However, as the line rate is increased, the sample rate 1/TS of the digital circuits within the transmitter and receiver must also increase, in order to maintain the T/2 sampling needed to avoid aliasing.
It will be appreciated that increased sample rates imply that the power consumption of the receiver must necessarily also increase, as will the heat generated by the circuits during run-time. This can impose an effective “thermal barrier” to increasing the line rate, as higher temperatures degrade system reliability. In addition, higher sampling rates are more difficult to implement in any practical integrated circuit (such as an Application Specific Integrated Circuit, ASIC, or a Field Programmable Gate Array, FPGA) due to sampling time jitter and limited bandwidth of the available circuit components.
Accordingly, techniques that enable reliable operation of digital signal processing systems at line rates above 10 Gbaud are highly desirable.